Optical signals received through conventional optical links are typically distorted by significant amounts of chromatic dispersion (CD) and polarization dependent impairments such as Polarization Mode Dispersion (PMD), polarization angle changes and polarization dependent loss (PDL). Chromatic dispersion (CD) on the order of 30,000 ps/nm, and polarization rotation transients at rates of 105 Hz are commonly encountered. Various methods and systems intended to address some of these limitations are known in the art.
FIG. 1 schematically illustrates a representative coherent optical receiver capable of implementing the methods of Applicant's U.S. Pat. No. 7,555,227 issued Jun. 30, 2009 and entitled “Polarization Compensation In A Coherent Optical Receiver”; U.S. Pat. No. 7,606,498 issued Oct. 20, 2009 and entitled “Carrier Recovery In A Coherent Optical Receiver”; and U.S. Pat. No. 7,636,525 issued Dec. 22, 2009 and entitled “Signal Acquisition In A Coherent Optical Receiver”, the content of all of which are hereby incorporated herein by reference.
As may be seen in FIG. 1, an inbound optical signal is received through an optical link 2, split into orthogonal received polarizations by a Polarization Beam Splitter 4, and then mixed with a Local Oscillator (LO) signal 6 by a conventional 90° optical hybrid 8. The composite optical signals emerging from the optical hybrid 8 are supplied to respective photodetectors 10, which generate corresponding analog electrical signals. The photodetector signals are sampled by respective Analog-to-Digital (A/D) converters 12 to yield raw multi-bit digital signals IX, QX and IY, QY corresponding to In-phase (I) and Quadrature (Q) components of each of the received polarizations.
Preferably, the raw multi-bit digital signals have resolution of n=5 or 6 bits which has been found to provides satisfactory performance at an acceptable cost. In the above-noted U.S. patent applications, the sample rate of the A/D converters 12 is selected to satisfy the Nyquist criterion for the highest anticipated symbol rate of the received optical signal. Thus, for example, in the case of an optical network link 2 having a line rate of 10 GBaud, the sample rate of the A/D converters 12 will be approximately 20 GHz.
From the A/D converter 12 block, the respective n-bit signals IX, QX and IY, QY of each received polarization are supplied to a respective dispersion compensator 14, which operates on the raw digital signals to at least partially compensate chromatic dispersion of the received optical signal. The dispersion compensators 14 may be configured to operate as described in Applicant's co-pending U.S. patent application Ser. No. 11/550,042 filed Oct. 17, 2006, and summarized below with reference to FIGS. 2a and 2b. 
As may be seen in FIG. 2a, each dispersion compensator (CD-COMP) 14 is provided as a high speed digital signal processor (or, equivalently, either an Application Specific Integrated Circuit, ASIC, or a Field Programmable Gate Array, FPGA) which is capable of implementing a variety of processing functions. In the illustrated embodiment, two substantially identical CD-COMPs 14 are provided, each of which is connected to receive and process raw In-phase and Quadrature digital signals of a respective received polarization. For simplicity only the X-polarization CD-COMP 14x is illustrated in FIG. 2a, it being understood that the Y-polarization CD-COMP 14y will be substantially identical.
In the embodiment of FIG. 2a, the CD-COMP 14 generally comprises a pipelined series of functional blocks, including a deserializer 24, a Fast Fourier Transform (FFT) filter 26, a frequency domain processor (FDP) 28 and an Inverse Fast Fourier Transform (IFFT) filter 30.
The deserializer 24 operates to accumulate successive n-bit words of the In-phase and Quadrature digital signals IX and QX from the X-polarization A/D converters 12IX and 12QX during a predetermined clock period. The accumulated n-bit words are then latched into the FFT 26 as a parallel input vector {rIX+jrQX}. Preferably, each of the real and imaginary components of the parallel vector {rIX+jrQX} have the same resolution (n=5 or 6 bits, for example) as the raw digital signals. In general, the width (m), in words, of the input vector {rIX+jrQX} is selected to be half the width (M) of the FFT 26. In some embodiments, the FFT 26 has a width of M=256 taps, which implies an input vector width of m=128 complex values. However, a different FFT width may be selected, as desired. In practice, the FFT width is selected based on a compromise between circuit size and the amount of dispersion compensation desired.
The input vector {rIX+jrQX} is augmented with a null vector {0, 0, 0, . . . 0} 32 which provides a zero data fill to the remaining input taps of the FFT 26.
The FFT filter 26 performs a conventional FFT operation to generate an array {RAX} representing the frequency domain spectrum of the input vector {rIX+jrQX}. The FDP 28 can then implement any of a variety of frequency domain processing functions, as will be described in greater detail below, to yield a modified array {VAX}, which is supplied to the IFFT filter 30.
The IFFT filter 30 performs a conventional Inverse Fast Fourier Transform operation to yield time domain data 34, in the form of a complex valued vector having a width equal to the IFFT 30, which, in the illustrated embodiment is M taps. In the embodiment of FIG. 2a, the IFFT output data 34 is divided into two blocks {v0X}, and {v1X}, of which {v1X} is delayed by one clock cycle (at 36) and added to {v0X} (at 38) to yield the CD-COMP output 16 in the form of a complex valued vector {vIX+jvQX} encompassing m(=128) complex values.
In the system of FIGS. 2a and 2b, the FDP 28 implements a transpose-and-add function, along with dispersion compensation. In general, the transpose-and-add function operates to add the FFT output vector {RAX} to a transposed version of itself {RXA}, with respective different compensation vectors. Implementing the transpose-and-add operation between the complex FFT and IFFT filters has the effect of emulating a pair of parallel real-FFT and IFFT functions through the CD-COMP 14, without requiring the additional circuits needed for parallel real FFT and IFFT filters. The transpose-and-add function can be conveniently implemented in hardware, by providing a pair of parallel paths between the FFT output and a vector addition block 40. One of these paths may be referred to as a direct path 42, in which the tap-order of the FFT output {RAX} is retained. The other path, which may be referred to as a transpose path 44, includes a transposition block 46 which operates to reverse the tap-order of the FFT output, upstream of the vector addition block 40. In this respect, it will be recognised that the transposition block 46 can be readily implemented in hardware, which provides an advantage in that the transposition step does not incur a significant propagation delay penalty.
Preferably, the direct and transpose paths 42 and 44 are provided with a respective multiplication block 48, which enables various filter functions to be implemented by the FDP 28. For example, in the embodiment of FIG. 2b, a pair of compensation vectors {C0X} and {CTX} 50 are applied to the direct and transpose paths, 42 and 44 respectively. Each of the compensation vectors {C0X} and {CTX} is composed of a respective set of coefficients which are calculated to apply a desired function, in the frequency-domain, to the digital signals. For example, {C0X} and {CTX} may be calculated to apply a first-order dispersive function to at least partially compensate chromatic dispersion of the optical link. {C0X} and {CTX} may also incorporate a transform of a differential delay function, so as to compensate residual sample phase errors in the I and Q digital signals. When both of these functions are implemented by the compensation vectors {C0X} and {CTX}, the CD-COMP output 16 will represent a dispersion-compensated and phase-error corrected version of the raw IX and QX digital signals received from the A/D converters 12.
Returning to FIG. 1, the dispersion-compensated digital signals 16 appearing at the output of the dispersion compensators 14 are then supplied to a polarization compensator 18 which operates to compensate polarization effects, and thereby de-convolve transmitted symbols from the complex signals 16 output from the dispersion compensators 14. If desired, the polarization compensator 18 may operate as described in Applicant's U.S. Pat. No. 7,555,227 issued Jun. 30, 2009 and U.S. Pat. No. 7,606,498 issued Oct. 20, 2009. The output of the polarization compensator 18 is a pair of multi-bit estimates X′(n) and Y′(n), 20 of the symbols encoded on each transmitted polarization. The symbol estimates X′(n), Y′(n) appearing at the output of the polarization compensator 18 are then supplied to a carrier recovery block 22 for LO frequency control, symbol detection and data recovery, such as described in Applicant's U.S. Pat. No. 7,606,498 issued Oct. 20, 2009.
In the above described system, the dispersion compensators 14 operates across a large number of successive samples (e.g. 128 samples), which permits compensation of relatively severe chromatic dispersion, but at a cost of a relatively slow response to changing dispersion. This slow response is acceptable, because of the known slow rate of change of dispersion in real-world optical links. The polarization compensator 18, in contrast, is comparatively very narrow (e.g. on the order of about 5 samples), to enable a rapid update frequency, which is necessary to track observed high-speed polarization transients.
The above-described system provides reliable signal acquisition, compensation of dispersion and polarization effects, carrier recovery and data recovery even in the presence of moderate-to-severe optical impairments. This, in turn, enables the deployment of a coherent optical receiver in real-world optical networks, with highly attractive signal reach and line rate characteristics. For example, a receiver implementing the above methods has demonstrated a signal reach of 1500 km at a line rate of 10 Gbaud (i.e. 109 symbols/second). It is noteworthy that this performance has been measured with real-time continuous processing, not just burst data acquisition followed by off-line processing or simulation. The system described above with reference to FIGS. 1 and 2 is the only coherent optical receiver known to the applicants to have achieved such real-time performance.
With increasing demand for link band-width, it would be desirable to increase the line rate beyond 10 Gbaud. For example, lines rates of 35 GBaud and higher have been proposed. However, as the symbol rate is increased, the amount of distortion compensation that is required in order to obtain the same signal reach also increases. For example, the required amount of dispersion compensation increases proportional to the square of the symbol rate, while the required amount of compensation for polarization effects increases proportional to the symbol rate. These increases in distortion compensation can be met, using the system described above, but at a cost of increased size and/or complexity of the dispersion and polarization compensation blocks.
At the same time, increasing the line rate also necessitates an increase in the sample rate of the A/D converters and downstream digital circuits, in order to maintain Nyquist sampling.
It will be appreciated that both increased circuit size and increased sample rate imply that the power consumption of the receiver must necessarily also increase, as will the heat generated by the circuits during run-time. This can impose an effective “thermal barrier” to increasing the line rate, as higher temperatures degrade system reliability.
Accordingly, methods and techniques that enable reliable operation of a coherent optical receiver at line rates above 10 Gbaud are highly desirable.